Method and apparatus for processing transmit power control (TPC) commands in a wideband CDMA (WCDMA) network

ABSTRACT

Method and apparatus for processing transmit power control (TPC) commands in a wideband CDMA (WCDMA) network are disclosed and may include calculating a signal-to-noise ratio (SNR) of a downlink dedicated physical channel (DPCH) based on a plurality of transmit power control (TPC) bits received via the downlink DPCH. A value of at least one of said plurality of TPC bits is not known when said at least one of said plurality of TPC bits is received. Transmit power for at least one uplink communication path may be adjusted based on the calculated SNR of the downlink dedicated physical channel. At least one reliability weight value may be calculated for at least a portion of the received TCP bits, based on the calculated SNR.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This application makes reference to:

U.S. application Ser. No. ______ (Attorney Docket No. 17001US01) filedon even date herewith;

U.S. application Ser. No. ______ (Attorney Docket No. 16999US01) filedon even date herewith; and

U.S. application Ser. No. ______ (Attorney Docket No. 17000US01) filedon even date herewith.

Each of the above state applications is hereby incorporated herein byreference in its entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to wireless communication.More specifically, certain embodiments of the invention relate to amethod and apparatus for processing transmit power control (TPC)commands in a wideband CDMA (WCDMA) network.

BACKGROUND OF THE INVENTION

Mobile communications has changed the way people communicate and mobilephones have been transformed from a luxury item to an essential part ofevery day life. The use of mobile phones is today dictated by socialsituations, rather than hampered by location or technology. While voiceconnections fulfill the basic need to communicate, and mobile voiceconnections continue to filter even further into the fabric of every daylife, the mobile Internet is the next step in the mobile communicationrevolution. The mobile Internet is poised to become a common source ofeveryday information, and easy, versatile mobile access to this datawill be taken for granted.

Third generation (3G) cellular networks have been specifically designedto fulfill these future demands of the mobile Internet. As theseservices grow in popularity and usage, factors such as cost efficientoptimization of network capacity and quality of service (QoS) willbecome even more essential to cellular operators than it is today. Thesefactors may be achieved with careful network planning and operation,improvements in transmission methods, and advances in receivertechniques. To this end, carriers need technologies that will allow themto increase downlink throughput and, in turn, offer advanced QoScapabilities and speeds that rival those delivered by cable modem and/orDSL service providers. In this regard, networks based on wideband CDMA(WCDMA) technology may make the delivery of data to end users a morefeasible option for today's wireless carriers.

In the case of a WCDMA downlink, multiple access interference (MAI) mayresult from inter-cell and intracell interference. The signals fromneighboring base stations compose intercell interference, which ischaracterized by scrambling codes, channels and angles of arrivalsdifferent from the desired base station signal. Spatial equalization maybe utilized to suppress inter-cell interference. In a synchronousdownlink application, employing orthogonal spreading codes, intra-cellinterference may be caused by multipath propagation. Due to the non-zerocross-correlation between spreading sequences with arbitrary timeshifts, there is interference between propagation paths afterdespreading, causing MAI. The level of intra-cell interference dependsstrongly on the channel response. In nearly flat fading channels, thephysical channels remain almost completely orthogonal and intra-cellinterference does not have any significant impact on the receiverperformance. Frequency selectivity is common for the channels in WCDMAnetworks.

Mobile networks allow users to access services while on the move,thereby giving end users freedom in terms of mobility. However, thisfreedom does bring uncertainties to mobile systems. The mobility of theend users causes dynamic variations both in the link quality and theinterference level, sometimes requiring that a particular user changeits serving base station. This process is known as handover (HO).Handover is the essential component for dealing with the mobility of endusers. It guarantees the continuity of the wireless services when themobile user moves across cellular boundaries.

WCDMA networks may allow a mobile handset to communicate with a multiplenumber of cell sites. This may take place, for example, for asoft-handoff from one cell site to another. Soft-handoffs may involvecell sites that use the same frequency bandwidth. On occasions, theremay be handoffs from one cell site to another where the two cell sitesuse different frequencies. In these cases, the mobile handset may needto tune to the frequency of the new cell site. Additional circuitry maybe required to handle communication over a second frequency of thesecond cell site while still using the first frequency for communicatingwith the first cell site. The additional circuitry may be an undesirableextra cost for the mobile handset. In addition, the mobile handset mayrequire different transmit power to establish and maintain acommunication link with the new cell site. In a handoff scenario, themobile handset may still be receiving a strong signal from the currentcell site and a weaker signal from the new cell site. In this regard,transmit power may have to be adjusted so that the handoff may beachieved and the mobile handset may begin to communicate with the newcell site.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or apparatus for processing transmit power control (TPC)commands in a wideband CDMA (WCDMA) network, substantially as shown inand/or described in connection with at least one of the figures, as setforth more completely in the claims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is an exemplary diagram illustrating a WCDMA handsetcommunicating with two WCDMA base stations, in accordance with anembodiment of the invention.

FIG. 1B is a block diagram of an exemplary radio frame format of adownlink dedicated physical channel (DPCH), in accordance with anembodiment of the invention.

FIG. 2 is a block diagram illustrating determination of reliabilityweights in a WCDMA network, in accordance with an embodiment of theinvention.

FIG. 3 is a flowchart illustrating exemplary steps for calculating asignal power estimate of the DPCH, in accordance with an embodiment ofthe invention.

FIG. 4 is a flowchart illustrating exemplary steps for calculating anoise power estimate of the DPCH, in accordance with an embodiment ofthe invention.

FIG. 5 is a block diagram of a system for weighted combination ofmultiple TPC commands, in accordance with an embodiment of theinvention.

FIG. 6 is a flowchart illustrating exemplary steps for determining atotal TPC command in a WCDMA network, in accordance with an embodimentof the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method andapparatus for processing transmit power control (TPC) commands in awideband CDMA (WCDMA) network. Aspects of the invention may comprisecalculating a signal-to-noise ratio (SNR) of a downlink dedicatedphysical channel (DPCH) based on a plurality of transmit power control(TPC) bits received via the downlink DPCH. A value of at least one ofsaid plurality of TPC bits may not be known when said at least one ofsaid plurality of TPC bits is received. Transmit power for at least oneuplink communication path may be adjusted based on the calculated SNR ofthe downlink dedicated physical channel. At least one reliability weightvalue may be calculated for at least a portion of the received TCP bits,based on the calculated SNR. A total TPC command may be generated forthe at least one uplink communication path based on the plurality ofreceived TPC bits and the calculated at least one reliability weightvalue. A selected reliability weight value may be discarded from thecalculation of the total TPC command, if the selected reliability weightvalue is higher than a threshold value. The transmit power may beadjusted for the uplink communication channel based on the calculatedtotal TPC command.

In accordance with an embodiment of the invention, methods forprocessing transmit power control (TPC) commands disclosed herein mayapply to diversity and non-diversity wireless systems. Diversitywireless systems may comprise space-time transmit diversity (STTD),closed loop 1 (CL1), and closed loop 2 (CL2) wireless systems.

In one embodiment of the invention, user equipment (UE) may be enabledto receive TPC commands transmitted on a downlink DPCH from one or moreradio links. The received TPC commands may be combined in a weightedfashion, and a final TPC decision may be generated depending on whetherto increase or decrease the user equipment transmit power. In thisregard, a reliability factor may be determined for each of the receivedTPC commands based on a signal-to-noise ratio (SNR) measurement. Thereliability factor may then be used to compute a weighted sum of themultiple received TPC commands, thereby yielding an accumulated TPCcommand. The sign of the accumulated TPC command may be used todetermine whether to increase, decrease or maintain the transmit power.

Uplink power control (PC) is of paramount importance for CDMA-basedsystems because the capacity of such a system is a function of theinterference level. The power transmitted by all active user equipments(UE) within a network may be controlled to limit interference levels andalleviate well-known problems such as the “near-far” effect. If there ismore than one user active, the transmitted power of non-reference usersis suppressed by a factor dependent on the partial cross-correlationbetween the code of the reference user and the code of the non-referenceuser. However, when a non-reference user is closer to the receiver thanthe reference user, it is possible that the interference caused by thisnon-reference user has more power than the reference user also referredto as the “near-far” effect.

There are two types of power-control techniques. Open-loop power-controlwhere each user equipment measures its received signal power and adjustsits transmit power accordingly and closed-loop power-control where anactive radio link (RL) measures the received signal power from all userequipments and simultaneously commands the individual user equipments toraise or lower their transmit uplink power such that the receivedsignal-to-noise ratio (SNR) from all user equipments at the radio linksis the same.

FIG. 1A is an exemplary diagram illustrating a WCDMA handsetcommunicating with two WCDMA base stations, in accordance with anembodiment of the invention. Referring to FIG. 1A, there is shown amobile handset or user equipment 120, a plurality of base stations BS122 and BS 124, and a plurality of radio links (RL), RL₁ and RL₂coupling the user equipment 120 with the base stations BS 122 and BS124, respectively. The user equipment 120 may comprise a processor 142,a memory 144, and a radio 146.

The processor 142 may communicate and/or control a plurality of bitsto/from the base stations BS 122 and BS 124. The memory 144 may comprisesuitable logic, circuitry, and/or code that may store data and/orcontrol information. The radio 146 may comprise transmit circuitryand/or receive circuitry that may be enabled to calculate asignal-to-noise ratio (SNR) of a downlink dedicated physical channel(DPCH) based on a plurality of transmit power control (TPC) bitsreceived via the downlink dedicated physical channel (DPCH), where theplurality of TPC bits may not be known when they are received. The radiolinks that belong to the same radio link set may broadcast the samevalues of transmit power control (TPC) bits. The radio links that belongto different radio link sets may broadcast different TPC bits. The userequipment 120 may receive TPC bits via multiple radio links, forexample, RL₁ and RL₂ simultaneously. In a handover situation, the userequipment 120 may simultaneously receive signals from multiple radiolink sets.

The WCDMA specification defines the physical random access channel(PRACH) for mobile phone uplinks and the acquisition indicator channel(AICH) for BTS downlinks. Communication is established when the userequipment 120 completes its search for a base station, for example, BS122 and synchronizes its PRACH uplink signal with the BTS AICH downlinksignal. When operating properly, the base station recognizes a PRACHpreamble from the user equipment 120 and responds with an AICH toestablish a communication link. The user equipment 120 may use the PRACHto transmit its setting of its open loop power control to the basestation 122. Incorrect data in the PRACH preamble or problems with thesignal quality may cause missed connections, disrupt the capacity of thecell or prevent response from the base station 122.

FIG. 1B is a block diagram of an exemplary radio frame format of adownlink dedicated physical channel (DPCH), in accordance with anembodiment of the invention. Referring to FIG. 1B, there is shown aradio frame format 102, with a time period T_(f) equal to 10 ms, forexample. The radio frame 102 may comprise a plurality of slots, forexample, 15 slots. Each of the slots in the radio frame 102, forexample, slot # 104 may comprise a plurality of dedicated physical datachannels (DPDCH) and a plurality of dedicated physical control channels(DPCCH). The time period of each slot in the radio frame 102, forexample, time period of slot # i may be equal to 10*2^(k) bits, wherek=0 . . . 7, for example.

The DPDCH is a type of downlink channel, which may be represented as anI/Q code multiplexed within each radio frame 102. The downlink DPDCH maybe utilized to carry data, for example, data 1 154 comprising N_(data1)bits and data 2 160 comprising N_(data2) bits. There may be zero, one,or a plurality of downlink dedicated physical data channels on eachradio link.

The DPCCH is a type of downlink channel, which may be represented as anI/Q code multiplexed within each radio frame 102. The downlink DPCCH maybe utilized to carry control information generated at the physicallayer. The control information may comprise a transmit power control(TPC) block 156 comprising N_(TPC) bits per slot, a transport formatcombination indicator (TFCI) block 158 comprising N_(TFCI) bits per slotand a pilot block 162 comprising N_(pilot) bits per slot.

Unlike the pilot bits 162 which are known a priori, that is, they areknown when received by a receiver, the TPC bits 156 may be known orunknown when they are received. The term “a priori” means “formed orconceived beforehand.” The phrase “not known” means that when some orall of the TPC bits are received at the receiver, the receiver cannotdetermine their actual values, and may need to determine the quality ofthe channel in order to determine whether the TPC bits are valid or not.Accordingly, various embodiments of the invention utilize channelquality to determine whether the TPC bits are valid or invalid.Therefore, conventional methods of computing a signal-to-noise ratio(SNR) metric based on multiplying the received signal by an a knownsequence may not be used here.

In an embodiment of the invention, the quality of the downlink controlchannel transmitted with the downlink dedicated physical channel (DPCH)may be determined. Within one downlink DPCH, dedicated data may betransmitted in time-multiplex manner with control information. Thecontrol information may comprise pilot bits, transport formatcombination indicator (TFCI) bits and transmit power control (TPC) bits.

The user equipment 120 may be enabled to estimate the quality ofreception of the TPC bits. The user equipment 120 may be, for example, ahandheld phone or a wireless card in a laptop computer, for example. Ifthe TPC bits are received under reliable channel conditions, they may bedemodulated correctly by the user equipment 120, which in turn maydetect correctly the power control commands sent down by the servingradio link, and adjust its transmit power appropriately, therebyavoiding interference. On the other hand, if the TPC bits are receivedunder poor channel conditions, the TPC commands may be decodedincorrectly by the user equipment 120, which in turn may be transmittinginappropriate transmit power levels, creating undesirable interferenceand limiting the system capacity.

In another embodiment of the invention, in instances when multiple RLsets are active, such as RL1 and RL2, multiple TPC commands may bereceived at the user equipment 120. The TPC commands derived from RL1and RL2 may comprise TPC bits, such as TPC bits 156. In addition, thereceived TPC bits from the multiple RL sets may be combined to determinea final TPC command for the user equipment 120. The final TPC commandmay be used by the user equipment 120 to make a decision as to whetherto increase or decrease its transmit power by a determined step size.

Since some TPC commands may be received by the user equipment 120 underbetter channel conditions than others, a different weight value may beassigned to each TPC command in a radio link set. In this regard, areliability factor may be determined for each of the one or more TPCcommands received by the user equipment 120 based on a signal-to-noiseratio measurement, for example. The reliability factor may be used tocompute a weighted sum of the multiple received TPC commands, resultingin the accumulated final TPC command. In addition, the reliabilityfactor of each received TCP command may be compared to a thresholdvalue. If the reliability factor for a particular received TCP commandis lower than the threshold value, the reliability factor and the TCPcommand may not be used in the calculation of the final TCP command Thesign of the final TPC command may be used to determine whether to stepup or down the transmit power of the user equipment 120.

FIG. 2 is a block diagram illustrating determination of reliabilityweights in a WCDMA network, in accordance with an embodiment of theinvention. Referring to FIG. 2, there is shown a plurality of TPCextraction fingers for a given radio link (RL), for example, TPCextraction finger i 202 through TPC extraction finger j 204, a pluralityof summing blocks 206, 208, 210, 214, 216, 226 and 234, a plurality ofsquaring blocks 212 and 228, a multiplier 218, a plurality of dividerblocks 220 and 230, a plurality of averaging blocks 222 and 232 and areliability weight generator block 224.

The signal-to-noise ratio (SNR), or equivalently the signal and noisepower components of the TPC command received from a given radio linkset, may be computed. A receiver technique that uses several basebandcorrelators to individually process several signal multipath components,for example, a rake receiver may be utilized. The correlator outputsalso known as fingers may be combined to achieve improved communicationsreliability and performance.

U.S. application Ser. No. 11/173,871 (Attorney Docket No. 16202US02)filed Jun. 30, 2005, provides a detailed description of a rake receiver,and is hereby incorporated herein by reference in its entirety.

In a multipath-fading environment, a receiver structure may assignfingers to the multiple received paths, for example, TPC extractionfinger i 202 and TPC extraction finger j 204. Those fingers belonging tothe same radio link (RL) set may be summed by the summing block 206 togenerate TPC_I_finger_sum (k) and TPC_Q_finger_sum (k) where k is indexof the RL set.

For signal power, the value of the TPC bits may not be known a prioribut all TPC bits received within a slot may have the same value.Therefore, by adding the I and Q components, the signal portion may additself coherently, while the noise may add itself incoherently. Thiseffect in a noise reduction and the signal power may be extracted. Thei-th received TPC bit at a given slot and finger j may be expressed as:$\begin{matrix}{{TPC\_ bit}_{ij} = {{\sqrt{\frac{S_{TPC}}{2}}s_{b_{i}}{h_{j}}^{2}} + {\underset{{or}\quad{imag}}{real}\quad\left( {h_{j}^{*}n_{ij}} \right)}}} & (1.)\end{matrix}$where S_(TPC) may be the signal power, s_(bi) may be the value of theTPC bit and may be either + or −1, h_(j) may be the complex channel gainat finger j and n_(ij) may be a complex random variable representing thenoise component of variance denoted by$I_{oc} + {I_{or}{\sum\limits_{k \neq j}{{h_{k}}^{2}.}}}$I_(or) may be the total transmit power spectral density of the downlinksignal at the base station antenna connector. I_(oc) may be the powerspectral density of a band limited white noise source (simulatinginterference from cells) as measured at the UE antenna connector.The fingers corresponding to the radio links belonging to the same RLset together may be summed according to the following equation:$\begin{matrix}{{TPC\_ bit}_{i} = {{\sqrt{\frac{S_{TPC}}{2}}s_{b_{i}}{\sum\limits_{j}{h_{j}}^{2}}} + {\sum\limits_{j}{\underset{{or}\quad{imag}}{real}\quad\left( {h_{j}^{*}n_{ij}} \right)}}}} & (2.)\end{matrix}$The TPC command may be the sum of the set of received TPC bits within aslot. Depending on the slot format, the number of TPC bits per slot,denoted by num_tpc may change. $\begin{matrix}{{TPC\_ cmd} = {{{num\_ tpc}\sqrt{\frac{S_{TPC}}{2}}s_{b_{i}}{\sum\limits_{j}{h_{j}}^{2}}} + {\sum\limits_{i}^{{num\_ tpc}/2}{\sum\limits_{j}{{real}\quad\left( {h_{j}^{*}n_{ij}} \right)}}} + {{imag}\quad\left( {h_{j}^{*}n_{ij}} \right)}}} & (3.) \\{{{TPC\_ cmd}^{2} = {{{num\_ tpc}^{2}\quad\frac{S_{TPC}}{2}\quad\left( {\sum\limits_{j}{h_{j}}^{2}} \right)^{2}} + \left( {{\sum\limits_{i}^{{num\_ tpc}/2}{\sum\limits_{j}{{real}\quad\left( {h_{j}^{*}\quad n_{ij}} \right)}}} + {{imag}\quad\left( {h_{j}^{*}\quad n_{ij}} \right)}} \right)^{2}}}{{{where}\quad{E\left\lbrack \left( {{\sum\limits_{i}^{{num\_ tpc}/2}{\sum\limits_{j}{{real}\quad\left( {h_{j}^{*}\quad n_{ij}} \right)}}} + {{imag}\quad\left( {h_{j}^{*}\quad n_{ij}} \right)}} \right)^{2} \right\rbrack}} = {\frac{num\_ tpc}{2}{\sum\limits_{j}{{h_{j}}^{2}{\left( {I_{oc} + {I_{or}{\sum\limits_{k \neq j}{h_{k}}^{2}}}} \right).}}}}}} & (4.)\end{matrix}$The SNR of TPC command to be estimated may be: $\begin{matrix}{{SNR}_{TPC\_ cmd} = {{num\_ tpc}\quad\frac{{S_{TPC}\left( {\sum\limits_{j}{h_{j}}^{2}} \right)}^{2}}{\sum\limits_{j}{{h_{j}}^{2}\left( {I_{oc} + {I_{or}{\sum\limits_{k \neq j}{h_{k}}^{2}}}} \right)}}}} & (5.)\end{matrix}$

The TPC bits may be received on I and Q components, composing a symbol.For example, if the total number of bits within a slot is equal to 2,TPC_bit₁ may be received on the I component as TPC_(I1), and TPC_bit₂may be received on the Q component as TPC_(Q1). If the total number ofbits within a slot may be equal to num_tpc, there may be num_tpc/2 Icomponents and num_tpc/2 Q components.

The TPC bits (I and Q) may be summed by summing blocks 210 and 226 togenerate TPC_sum (k), where num_tpc may be the number of TPC bits perslot and k is the index of a given Radio Link set. The generated sumTPC_sum (k) may be squared by the squaring block 228 to generateTPC_sum_sqr (k) and a new estimate may be obtained once per slot. Thegenerated TPC_sum_sqr (k) may be divided by the number of TPC bits bythe divider block 230 to generate TPC_sum_sqr_norm (k) according to thefollowing equation:TPC_sum_(—) sqr _(—) norm(k)=TPC_sum_(—) sqr(k)/num _(—) tpcThe generated norm TPC_sum_sqr_norm (k) may be averaged over a giventime window by the averaging block 232 to generate TPC_sum_sqr_avg (k).An integrate-and-dump method, or an IIR filter may be utilized to carryout the averaging operation, for example.

In an embodiment of the invention, the signal power Ŝ_(tpc) may becomputed according to the following equations: $\begin{matrix}{{{TPC\_ sum}{\_ sqr}{\_ norm}} = {\frac{1}{num\_ tpc}\left( {{\sum\limits_{i = 1}^{{num\_ tpc}/2}{TPC}_{li}} + {TPC}_{Qi}} \right)^{2}}} & (6.)\end{matrix}$In additive white gaussian noise (AWGN), $\begin{matrix}{{\hat{S}}_{tpc} = {{E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack} = {{\frac{S_{tpc}}{2}{num\_ tpc}} + \frac{I_{oc}}{2}}}} & (7.)\end{matrix}$In flat fading, $\begin{matrix}{{\hat{S}}_{tpc} = {{E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack} = {{\frac{S_{tpc}}{2}{h}^{4}{num\_ tpc}} + {\frac{I_{oc}}{2}{h}^{2}}}}} & (8.)\end{matrix}$where h may be the complex channel gain at the finger.In space time transmit diversity (STTD) flat fading, $\begin{matrix}\begin{matrix}{{\hat{S}}_{tpc} = {E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack}} \\{= {{\frac{S_{tpc}}{4}\left( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} \right)^{2}{num\_ tpc}} + {\frac{I_{oc}}{2}\left( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} \right)}}}\end{matrix} & (9.)\end{matrix}$where h_(m) is the complex channel gain corresponding to transmitantenna m in the base station.In closed loop 1 (CL1) flat fading, $\begin{matrix}\begin{matrix}{{\hat{S}}_{tpc} = {E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack}} \\{= {{\frac{S_{tpc}}{4}{{h_{1} + {w\quad h_{2}}}}^{4}{num\_ tpc}} + {\frac{I_{oc}}{2}{{h_{1} + {w\quad h_{2}}}}^{2}}}}\end{matrix} & (10.)\end{matrix}$where h₁ and h₂ are the complex channel gains from the base stationtransmit antennas 1 and 2 and w is a weight.In closed loop 2 (CL2) fading, $\begin{matrix}\begin{matrix}{{\hat{S}}_{tpc} = {E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack}} \\{= {{\frac{S_{tpc}}{4}{{{w_{1}h_{1}} + {w_{2}\quad h_{2}}}}^{4}{num\_ tpc}} + {\frac{I_{oc}}{2}{{{w_{1}h_{1}} + {w_{2}\quad h_{2}}}}^{2}}}}\end{matrix} & (11.)\end{matrix}$where h₁ and h₂ are the complex channel gains from the base stationtransmit antennas 1 and 2 and w₁ and w₂ are weights.

In another embodiment of the invention, the signal power estimate may befurther improved by computing Stpc_avg (k) using the summer block 234according to the following equation:Stpc _(—) avg(k)=TPC_sum_(—) sqr _(—) avg(k)−Ntpc _(—) avg(k)  (12.)where Ntpc_avg (k) may be the noise power estimate.

An alternative embodiment of the invention may implement a differentcalculation of the signal power by squaring all TPC bits (I and Q). Thesquared TPC bits may be summed to generate TPC_sum_sqr (k) and a newestimate may be obtained once per slot. The generated TPC_sum_sqr (k)may be divided by the number of TPC bits to generate TPC_sum_sqr_norm(k) according to the following equation:TPC_sum_(—) sqr _(—) norm(k)=TPC_sum_(—) sqr(k)/num _(—) tpc  (13.)The generated norm TPC_sum_sqr (k) may be averaged over a given timewindow to generate TPC_sum_sqr_avg (k).

In an embodiment of the invention, the signal power Ŝ_(tpc) may becomputed according to the following equations: $\begin{matrix}{{{TPC\_ sum}{\_ sqr}{\_ norm}} = {{\frac{1}{num\_ tpc}{\sum\limits_{i = 1}^{{num\_ tpc}/2}{TPC}_{Ii}^{2}}} + {TPC}_{Qi}^{2}}} & (14.) \\{{\hat{S}}_{tpc} = {{E\quad\left\lbrack {{TPC\_ sum}{\_ sqr}{\_ norm}} \right\rbrack} = {\frac{S_{tpc}}{2} + \frac{I_{oc}}{2}}}} & (15.)\end{matrix}$

In another embodiment of the invention, the signal power estimate may befurther improved by computing Stpc_avg(k) according to the followingequation:Stpc _(—) avg(k)=TPC_sum_(—) sqr _(—) avg(k)−Ntpc _(—) avg(k)  (16.)and may be scaled by the average number of TPC bits over the averagingtime period according to the following equation:Stpc _(—) avg(k)=Stpc _(—) avg(k)*num _(—) tpc _(—) avg(k)  (17.)where num_tpc may vary from slot to slot.

For noise power, the value of the TPC bits may not be known a priori butall TPC bits received within a slot may have the same value. Therefore,by subtracting the I component from the Q component or vice-versa, thesignal portion may cancel itself out, leaving the residual noise.

In an embodiment of the invention, the noise power may be computed fromTPC bits only. The sign bit on both the I and Q components of the TPCsymbol may be the same. Therefore for each symbol,TPC _(I) −TPC _(Q) =n _(I) −n _(Q)  (18.)Since there are $\frac{num\_ tpc}{2}$symbols per slot, there may be $\frac{num\_ tpc}{2}$noise samples per slot. In AWGN, the noise power estimate may begenerated according to the following equation: $\begin{matrix}\begin{matrix}{{\hat{N}}_{tpc} = {E\left\lbrack {\sum\limits_{i = 1}^{{num\_ tpc}/2}\left( {{TPC}_{Ii} - {TPC}_{Qi}} \right)^{2}} \right\rbrack}} \\{= {E\left\lbrack {\sum\limits_{i = 1}^{{num\_ tpc}/2}\left( {n_{Ii} - n_{Qi}} \right)^{2}} \right\rbrack}} \\{= {{num\_ tpc} \cdot \frac{I_{oc}}{2}}}\end{matrix} & (19.)\end{matrix}$

In flat fading, the noise power estimate may be generated according tothe following equation: $\begin{matrix}{{\hat{N}}_{tpc} = {{h}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (20.)\end{matrix}$In STTD flat fading, the noise power estimate may be generated accordingto the following equation: $\begin{matrix}{{\hat{N}}_{tpc} = {\left( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} \right){{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (21.)\end{matrix}$In CL1 flat fading, the noise power estimate may be generated accordingto the following equation: $\begin{matrix}{{\hat{N}}_{tpc} = {{{h_{1} + {w\quad h_{2}}}}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (22.)\end{matrix}$In CL2 fading, the noise power estimate may be generated according tothe following equation: $\begin{matrix}{{\hat{N}}_{tpc} = {{{{w_{1}h_{1}} + {w_{2}\quad h_{2}}}}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (23.)\end{matrix}$

The TPC bits may be subtracted from each other (I−Q) by the summingblock 208. The subtracted TPC bits may be squared by the squaring block212 to generate TPC_sqr_diff (k). The squared difference TPC_sqr_diff(k) may be summed by the summing block 214 over the number of TPCsymbols, where the number of symbols per slot may be equal to num_tpc/2to generate Ntpc (k) and a new estimate may be obtained once per slot.The sum Ntpc (k) may be divided by the number of TPC bits by the dividerblock 220 to generate Ntpc_norm (k) according to the following equation:Ntpc _(—) norm(k)=Ntpc(k)/num _(—) tpc  (24.)The generated norm Ntpc_norm (k) may be averaged by the averaging block222 over a given time window to generate Ntpc_avg (k).

An alternative embodiment of the invention may improve the accuracy onthe noise power estimate. The noise power may be computed based on theTPC bits received within a slot. For slot formats with a small number ofTPC bits per slot, for example, 2 TPC bits per slot, the variance of thenoise power estimate may be quite large. This embodiment improves thenoise estimate by augmenting the noise estimated from TPC bits by othersources of noise estimates. By adding extra samples of noise estimatesfor a given slot and averaging over the total number of noise samplesavailable, the variance of the estimate may be reduced or the estimatemay be more accurate.

In an exemplary embodiment of the invention, the noise estimate may beaugmented from the estimate obtained from the reception of the dedicatedpilot bits (block 162 on FIG. 1B), or the common pilot bits (CPICH). Ascaling factor denoted by A, between the outsourced noise power estimateNout and the noise power estimate from the TPC bits may be used, and theimproved noise estimate Ntpc_aug (k) may be computed using themultiplier 218 according to the following equation:Ntpc _(—) aug(k)=(Ntpc(k)+A*Nout(k))/2  (25.)A is a scaling factor that may be dependent upon the number of TPC bitsper slot.

In an embodiment of the invention, the noise power may be computed froma combination of TPC bits and pilot bits. In a non-diversity flat fadingcase, the soft value of each of the dedicated pilot bits at each slot onmay be obtained from the hardware and the i-th pilot symbol may berepresented by the following equation: $\begin{matrix}{z_{i} = {{\sqrt{\frac{S_{DED}}{2}}x_{i}{h}^{2}} + {n_{i}h^{*}}}} & (26.)\end{matrix}$The number of dedicated pilot bits per slot may be denoted by num_dedand all num_ded/2 dedicated pilot symbols may be stacked in a vectoraccording to the following equation: $\begin{matrix}{\underset{\_}{z} = {{\sqrt{\frac{S_{DED}}{2}}{h}^{2}\underset{\_}{x}} + {\underset{\_}{n}}^{\prime}}} & (27.)\end{matrix}$where n′ may be the post-combining noise of the power to be estimated.The pilot symbol sequence${\underset{\_}{x}}^{T} = \left\lbrack {x_{0},x_{1},x_{2},\ldots\quad,x_{\frac{\quad_{num\_ ded}}{2} - 1}} \right\rbrack$may be known a priori, and it may be possible to find an orthogonalsequence${\underset{\_}{y}}^{T} = \left\lbrack {y_{0},y_{1},y_{2},\ldots\quad,y_{\frac{\quad_{num\_ ded}}{2} - 1}} \right\rbrack$such thaty ^(H) x=0  (28.)Since the pilot symbols are comprised of −1s and 1s, the sequence in ymay be also comprised of −1s and 1s. Multiplying the received symbols zby y ^(H) involves a sign change manipulation on the received I and Qand results in the following equation:y ^(H) z=y ^(H) n′  (29.)The variance of n′ may be expressed by the following equation:$\begin{matrix}{{\sigma_{n^{\prime}}^{2} = {{{h}^{2}I_{oc}} = {E\left\lbrack {n_{i}^{\prime}n_{i}^{\prime*}} \right\rbrack}}},{i = 0},\ldots\quad,{\frac{num\_ ded}{2} - 1}} & (30.)\end{matrix}$If the orthogonal sequence y may be normalized such thaty ^(H) y=1  (31.)Then the variance of y ^(H) n′ may be expressed as: $\begin{matrix}{{E\left\lbrack {{\underset{\_}{y}}^{H}{\underset{\_}{n}}^{\prime}{\underset{\_}{n}}^{\prime\quad H}\underset{\_}{y}} \right\rbrack} = \sigma_{n^{\prime}}^{2}} & (32.)\end{matrix}$From equation (25.), the noise power from TPC bits may be expressed as:$\begin{matrix}{{\hat{N}}_{tpc} = {{h}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (33.)\end{matrix}$Total noise estimate may be expressed as: $\begin{matrix}{\hat{N} = {\left( {{\hat{N}}_{tpc} + {\frac{num\_ tpc}{2} \cdot \sigma_{n^{\prime}}^{2}}} \right)/2}} & (34.)\end{matrix}$

In the case of flat fading, STTD with the number of pilot bits>2, thesoft value of each dedicated pilot bits at each slot may be obtained.The i-th received dedicated pilot symbol for antenna 1 may be equal to:$\begin{matrix}{z_{1\quad i} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{x_{1\quad i}h_{1}} + {x_{2\quad i}h_{2}}} \right)h_{1}^{*}} + {n_{i}h_{1}^{*}}}} & (35.)\end{matrix}$Similarly for antenna 2, $\begin{matrix}{z_{2\quad i} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{x_{1\quad i}h_{1}} + {x_{2\quad i}h_{2}}} \right)h_{2}^{*}} + {n_{i}h_{2}^{*}}}} & (36.)\end{matrix}$All num_ded/2 dedicated pilot symbols may be stacked in a vectoraccording to the following equations: $\begin{matrix}{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{{\underset{\_}{x}}_{1}h_{1}} + {{\underset{\_}{x}}_{2}h_{2}}} \right)h_{1}^{*}} + {\underset{\_}{n}}_{1}^{\prime}}} & (37.) \\{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{{\underset{\_}{x}}_{1}{h_{1}}^{2}} + {{\underset{\_}{x}}_{2}h_{2}h_{1}^{*}}} \right)} + {\underset{\_}{n}}_{1}^{\prime}}} & (38.) \\{{{\underset{\_}{z}}_{1} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}}^{2} \\{h_{2}h_{1}^{*}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime}}},{and}} & (39.) \\{{\underset{\_}{z}}_{2} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}h_{2}^{*}} \\{h_{2}}^{2}\end{bmatrix}} + {{\underset{\_}{n}}_{2}^{\prime}.}}} & (40.)\end{matrix}$

The pilot symbol sequences x ₁ and x ₂ are known a priori and it may bepossible to find an orthogonal sequence y ^(T) such thaty ^(H) x ₁=0 and y ^(H) x ₂=0  (41.)y ^(H) z ₁ =y ^(H) n ₁′ and y ^(H) z ₂ =y ^(H) n ₂′  (42.)The variance of n₁′ may be $\begin{matrix}{{\sigma_{n_{1}^{\prime}}^{2} = {{{h_{1}}^{2}I_{oc}} = {E\left\lbrack {n_{1\quad i}^{\prime}n_{1\quad i}^{\prime*}} \right\rbrack}}},{i = 0},\ldots\quad,{\frac{num\_ ded}{2} - 1},{and}} & (43.) \\{\sigma_{n_{2}^{\prime}}^{2} = {{h_{2}}^{2}{I_{oc}.}}} & (44.)\end{matrix}$If the orthogonal sequence y may be normalized such thaty ^(H) y =1  (45.)Then the variance of y ^(H) n _(i)′ may be $\begin{matrix}{{{E\left\lbrack {{\underset{\_}{y}}^{H}{\underset{\_}{n}}_{i}^{\prime}{\underset{\_}{n}}_{i}^{\prime\quad H}\underset{\_}{y}} \right\rbrack} = \sigma_{n_{i}^{\prime}}^{2}},{i = 1},2} & (46.)\end{matrix}$In this regard, the noise power from the dedicated pilot bits may beobtained by the following equation: $\begin{matrix}{{{{{\underset{\_}{y}}^{H}{\underset{\_}{z}}_{1}}}^{2} + {{{\underset{\_}{y}}^{H}{\underset{\_}{z}}_{2}}}^{2}} = {{\sigma_{n_{1}^{\prime}}^{2} + \sigma_{n_{2}^{\prime}}^{2}} = {\left( {{h_{1}}^{2} + {h_{2}}^{2}} \right)I_{oc}}}} & (47.)\end{matrix}$From equation (32) the noise power from TPC bits may be $\begin{matrix}{{\hat{N}}_{tpc} = {\left( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} \right){{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (48.)\end{matrix}$Total noise estimate may be: $\begin{matrix}{\hat{N} = {\left( {{\hat{N}}_{tpc} + {\frac{num\_ tpc}{2} \cdot \left( {\sigma_{n_{1}^{\prime}}^{2} + \sigma_{n_{2}^{\prime}}^{2}} \right)}} \right)/2}} & (49.)\end{matrix}$

When the number of pilot bits=2, the 2 pilot bits broadcast by antenna 2precede the last two bits of the data2 field. The pilot bits may beSTTD-encoded with the data and, therefore, may need to be retrievedpost-STTD decoding. The hardware may be provisioned to extract pilotbits at the output of the combiner, post-STTD decoding. The pilot symbolobtained post-STTD decoding may be: $\begin{matrix}{{z = {{\sqrt{\frac{S_{DED}}{4}}x_{1}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}}}},} & (50.)\end{matrix}$where x₁ may be the known pilot symbol sent from antenna 1 and$\begin{matrix}{{E\left\lbrack \left( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} \right)^{2} \right\rbrack} = {\left( {\sum\limits_{m = 1}^{2}{h_{m}}^{2}} \right){I_{oc}.}}} & \quad \\{{{pilot}\quad I} = {{{Re}\quad(z)} = {{\sqrt{\frac{S_{DED}}{4}}I_{seq}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {{Re}\left( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} \right)}}}} & (51.) \\{{{pilot}\quad Q} = {{{Im}\quad(z)} = {{\sqrt{\frac{S_{DED}}{4}}Q_{seq}{\sum\limits_{m = 1}^{2}{h_{m}}^{2}}} + {{Im}\left( {\sum\limits_{m = 1}^{2}{h_{m}^{*}n_{m}}} \right)}}}} & (52.)\end{matrix}$The hardware multiplies pilot I and pilot Q by I_(seq) and Q_(seq)respectively and generates the 2 bits. The noise power may be calculatedby the following equations: $\begin{matrix}{\sigma_{n}^{2} = \left( {{{pilot}\quad I} - {{pilot}\quad Q}} \right)^{2}} & (53.) \\{\sigma_{n}^{2} = {\left( {{h_{1}}^{2} + {h_{2}}^{2}} \right)I_{oc}}} & (54.)\end{matrix}$The total noise estimate may be expressed as: $\begin{matrix}{\hat{N} = {\left( {{\hat{N}}_{tpc} + {\frac{num\_ tpc}{2} \cdot \sigma_{n}^{2}}} \right)/2}} & (55.)\end{matrix}$

In the case of CL1 flat fading, the soft value of each dedicated pilotbits at each slot on a per-finger basis may be obtained from thehardware. $\begin{matrix}{{{\underset{\_}{z}}_{1} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}}^{2} \\{w\quad h_{1}^{*}h_{2}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime}}}{And}} & (56.) \\{{\underset{\_}{z}}_{2} = {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{h_{1}h_{2}^{*}} \\{w{h_{2}}^{2}}\end{bmatrix}} + {\underset{\_}{n}}_{2}^{\prime}}} & (57.)\end{matrix}$The weight w may be known in the firmware, $\begin{matrix}\begin{matrix}{\underset{\_}{z} = {{\underset{\_}{z}}_{1} + {w^{*}{\underset{\_}{z}}_{2}}}} \\{= {{{\sqrt{\frac{S_{DED}}{4}}\begin{bmatrix}{\underset{\_}{x}}_{1} & {\underset{\_}{x}}_{2}\end{bmatrix}}\begin{bmatrix}{{h_{1}}^{2} + {w^{*}h_{1}h_{2}^{*}}} \\{{w\quad h_{1}^{*}h_{2}} + {{w}^{2}{h_{2}}^{2}}}\end{bmatrix}} + {\underset{\_}{n}}_{1}^{\prime} + {w^{*}{\underset{\_}{n}}_{2}^{\prime}}}}\end{matrix} & (58.)\end{matrix}$Multiplying z by the orthogonal sequence y, $\begin{matrix}{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{\underset{\_}{y}}^{H}\left( {{\underset{\_}{n}}_{1}^{\prime} + {w^{*}{\underset{\_}{n}}_{2}^{\prime}}} \right)}} & (59.) \\{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{\underset{\_}{y}}^{H}\left( {\begin{bmatrix}{n_{0}h_{1}^{*}} \\\vdots \\{n_{\frac{num\_ ded}{2} - 1}h_{1}^{*}}\end{bmatrix} + {w^{*}\begin{bmatrix}{n_{0}h_{2}^{*}} \\\vdots \\{n_{\frac{num\_ ded}{2} - 1}h_{2}^{*}}\end{bmatrix}}} \right)}} & (60.) \\{{{\underset{\_}{y}}^{H}\underset{\_}{z}} = {{{\underset{\_}{y}}^{H}\left( \begin{bmatrix}{n_{0}\left( {h_{1}^{*} + {w^{*}h_{2}^{*}}} \right)} \\\vdots \\{n_{\frac{num\_ ded}{2} - 1}\left( {h_{1}^{*} + {w^{*}h_{2}^{*}}} \right)}\end{bmatrix} \right)} = {{\underset{\_}{y}}^{H}{\underset{\_}{n}}_{{cl}\quad 1}}}} & (61.)\end{matrix}$The variance of n _(cl1) may be $\begin{matrix}{{\sigma_{{\underset{\_}{n}}_{{cl}\quad 1}}^{2} = {{{{h_{1} + {w\quad h_{2}}}}^{2}I_{oc}} = {E\left\lbrack {n_{{cl}\quad 1\quad i}n_{{cl}\quad 1\quad i}^{*}} \right\rbrack}}},{i = 0},\ldots\quad,{\frac{num\_ ded}{2} - 1}} & (62.) \\{{{{\underset{\_}{y}}^{H}\underset{\_}{z}}}^{2} = \sigma_{{\underset{\_}{n}}_{{cl}\quad 1}}^{2}} & (63.)\end{matrix}$From equation (27.) the noise power from TPC bits may be $\begin{matrix}{{\hat{N}}_{tpc} = {{{h_{1} + {w\quad h_{2}}}}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (64.)\end{matrix}$Total noise estimate may be: $\begin{matrix}{\hat{N} = {\left( {{\hat{N}}_{tpc} + {\frac{num\_ tpc}{2} \cdot \sigma_{{\underset{\_}{n}}_{{cl}\quad 1}}^{2}}} \right)/2}} & (65.)\end{matrix}$

In the case of CL2 fading, the same pilot pattern may be used on boththe antennas. $\begin{matrix}{z_{1\quad i} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{w_{1}h_{1}} + {w_{2}h_{2}}} \right)x_{i}h_{1}^{*}} + {n_{i}h_{1}^{*}}}} & (66.)\end{matrix}$All num_ded/2 dedicated pilot symbols may be stacked in a vectoraccording to the following equations: $\begin{matrix}{{\underset{\_}{z}}_{1} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{w_{1}h_{1}} + {w_{2}h_{2}}} \right)\underset{\_}{x}\quad h_{1}^{*}} + {\underset{\_}{n}\quad h_{1}^{\prime}}}} & (67.) \\{{\underset{\_}{z}}_{2} = {{\sqrt{\frac{S_{DED}}{4}}\left( {{w_{1}h_{1}} + {w_{2}h_{2}}} \right)\underset{\_}{x}\quad h_{2}^{*}} + {\underset{\_}{n}\quad h_{2}^{\prime}}}} & (68.)\end{matrix}$The weights w₁ and w₂ are known in the firmware, $\begin{matrix}{\underset{\_}{z} = {{w_{1}^{*}{\underset{\_}{z}}_{1}} + {w_{2}^{*}{\underset{\_}{z}}_{2}}}} & (69.) \\{\underset{\_}{z} = {{\sqrt{\frac{S_{DED}}{4}}{{{w_{1}h_{1}} + {w_{2}h_{2}}}}^{2}\underset{\_}{x}} + {w_{1}^{*}{\underset{\_}{n}}_{1}^{\prime}} + {w_{2}^{*}{\underset{\_}{n}}_{2}^{\prime}}}} & (70.)\end{matrix}$Multiplying z by the orthogonal sequence y, $\begin{matrix}{{{{\underset{\_}{y}}^{H}\underset{\_}{z}}}^{2} = {\sigma_{{\underset{\_}{n}}_{{cl}\quad 2}}^{2} = {{{{w_{1}h_{1}} + {w_{2}h_{2}}}}^{2}I_{oc}}}} & (71.)\end{matrix}$From equation (34.) the noise power from TPC bits may be $\begin{matrix}{{\hat{N}}_{tpc} = {{{{w_{1}h_{1}} + {w_{2}\quad h_{2}}}}^{2}{{num\_ tpc} \cdot \frac{I_{oc}}{2}}}} & (72.)\end{matrix}$Total noise estimate may be: $\begin{matrix}{\hat{N} = {\left( {{\hat{N}}_{tpc} + {\frac{num\_ tpc}{2} \cdot \sigma_{{\underset{\_}{n}}_{{cl}\quad 2}}^{2}}} \right)/2}} & (73.)\end{matrix}$

The various embodiments of the invention described above may yield a TPCcommand signal and noise power estimate for each one of a plurality ofradio link sets. The signal and noise power estimate may be updatedperiodically, for example, at the rate of once per slot. In oneembodiment of the invention, a TPC command signal and noise powerestimate may be used to determine a reliability weight valuecorresponding to the received TPC command. A total or accumulated TPCcommand may then be determined based on the received TPC commands foreach one of the pluralities of radio link sets and the correspondingreliability weights for each of the TPC commands. Transmit power maythen be adjusted based on the determined total TPC command. For example,if the sign of total TPC command is negative, the transmit power may bedecreased, and if the sign of total TPC command is positive, thetransmit power may be increased.

FIG. 3 is a flowchart illustrating exemplary steps to calculate a signalpower estimate of the DPCH, in accordance with an embodiment of theinvention. Referring to FIGS. 2 and 3, exemplary steps may begin at step302. In step 304, the transmit power control (TPC) bits may be extractedfrom the slot received at each finger. In step 306, the TPC bitsextracted from all the fingers may be summed. In step 308, an Icomponent and a Q component may be generated from the summed TPC bits ofall the fingers. For signal power, the value of the TPC bits may not beknown a priori but all TPC bits received within a slot may have the samevalue. Therefore, by adding the I and Q components, the signal portionmay add itself coherently, while the noise may add itself incoherently.This effect in a noise reduction and the signal power may be extracted.In step 310, the TPC bits (I and Q) may be summed by the summing block210 to generate TPC_sum (k), where num_tpc is the number of TPC bits perslot. In step 312, the generated sum TPC_sum (k) may be squared by thesquaring block 228 to generate TPC_sum_sqr (k) and a new estimate may beobtained once per slot. In step 314, the generated TPC_sum_sqr (k) maybe divided by the number of TPC bits per slot, num_tpc, by the dividerblock 230 to generate TPC_sum_sqr_norm (k) according to the followingequation:TPC_sum_(—) sqr _(—) norm(k)=TPC_sum_(—) sqr(k)/num _(—) tpc

In step 316, the generated norm TPC_sum_sqr_norm (k) may be averaged bythe averaging block 232 over a given time window to generateTPC_sum_sqr_avg (k). In step 318, a signal power estimate of the TPCbits in the DPCH may be estimated. Control passes to end step 320.

FIG. 4 is a flowchart illustrating exemplary steps to calculate a noisepower estimate of the DPCH, in accordance with an embodiment of theinvention. Referring to FIGS. 2 and 4, exemplary steps may begin at step402. In step 404, the transmit power control (TPC) bits may be extractedfrom the slot received at each finger. In step 406, the TPC bitsextracted from all the fingers may be summed. In step 408, an Icomponent and a Q component may be generated from the summed TPC bits ofall the fingers. For noise power, the value of the TPC bits may not beknown a priori but all TPC bits received within a slot may have the samevalue. Therefore, by subtracting the I component from the Q component orvice-versa, the signal portion cancels itself out, leaving the residualnoise.

In step 410, the TPC bits may be subtracted from each other (I−Q) by thesumming block 208. In step 412, the subtracted TPC bits may be squaredby the squaring block 212 to generate TPC_sqr_diff (k). In step 414, thesquared difference TPC_sqr_diff (k) may be summed by the summing block214 over the number of TPC symbols, where the number of symbols per slotis equal to num_tpc/2 to generate Ntpc (k) and a new estimate may beobtained once per slot. In step 416, the sum Ntpc (k) may be divided bythe number of TPC bits by the divider block 220 to generate Ntpc_norm(k) according to the following equation:Ntpc _(—) norm(k)=Ntpc(k)/num _(—) tpcIn step 418, the generated norm Ntpc_norm (k) may be averaged by theaveraging block 222 over a given time window to generate Ntpc_avg (k).In step 420, the noise power estimate of the TPC bits in the DPCH may beestimated. Control passes to end step 422.

FIG. 5 is a block diagram of a system for weighted combination ofmultiple TPC commands, in accordance with an embodiment of theinvention. Referring to FIG. 5, the system 500 may comprise a pluralityof received TPC commands 502 a, . . . , 502 n, a plurality of signextraction blocks 504 a, . . . , 504 n, a plurality of multipliers 506a, . . . , 506 n, a plurality of zero multiplication blocks 505 a, . . ., 505 n, an adder 510, and a transmit power adjustment block 514. Thereceived TPC commands 502 a, . . . , 502 n may correspond to radio linksets 1, . . . , k, respectively. In this regard, a total of k receivedTPC commands may be used in the determination of a final or adjusted TPCcommand 512.

The sign extraction blocks 504 a, . . . , 504 n may comprise suitablecircuitry, logic, and/or code and may enable determination of the signof a corresponding TPC command. In this regard, the sign extractionblocks 504 a, . . . , 504 n may generate either (−1) or (+1) as a finalresult. The generated signs may be communicated to the correspondingmultipliers 506 a, . . . , 506 n. The multipliers 506 a, . . . , 506 nmay comprise suitable circuitry, logic, and/or code and may enablemultiplication of the received sign by a corresponding reliabilityweight value 508 a, . . . , 508 n.

In one embodiment of the invention, it may be determined whether each ofthe reliability weight values 508 a, . . . , 508 n is lower than thereliability_threshold. If a reliability weight value is lower than thereliability_threshold, the weighted sign value may be multiplied by zeroby a corresponding zero multiplication block from the plurality of zeromultiplication blocks 505 a, . . . , 505 n. In this regard, if thereliability weight value is lower than the reliability_threshold, thecorresponding weighted sign value may not be included in thedetermination of the final TPC command 512.

If the reliability weight value is higher than thereliability_threshold, the weighted sign values may be added by theadder 510 to generate the total TPC command 512. The transmit poweradjustment block 514 may comprise suitable circuitry, logic, and/or codeand may enable adjustment of the transmit power based on the determinedfinal TPC command 512. The final TPC command 512 may be used to adjustthe transmit power based on, for example, the sign of the final TPCcommand 512.

In one embodiment of the invention, the received TPC commands 502 a, . .. 502 n may belong to the same radio link (RL) set. Since radio linksbelonging to the same RL set transmit the same TPC command, the TPCcommands originating from radio links belonging to the same RL set maybe combined with equal weights. In this regard, the reliability weights508 a, . . . , 508 n may be the same, for example 1 or −1.

In another embodiment of the invention, the received TPC commands 502 a,. . . , 502 n may belong to different RL sets. For example, the receivedTPC commands 502 a, . . . , 502 n may belong to RL sets 1, . . . , K,respectively. In this regard, there may be one TPC command for each ofthe K RL sets, TPC_cmd(k), k=1, . . . K. The overall accumulated commandTPC_cmd 512 may be computed using the following exemplary pseudo code:Initialize the accumulated command to zero. Accum_cmd = 0 For (k=loopover RL sets) { Take sign of TPC_cmd(k) Accum_cmd + = (sign ofTPC_sum(k) ) * wk }where wk are the reliability weights 508 a, . . . , 508 n.

The value of Accum_cmd may correspond to the total TPC command 512. Thetransmit power adjustment block 514 may determine whether to increase ordecrease the transmit power based on the sign of Accum_cmd. For example,if the sign of Accum_cmd is negative, the transmit power may bedecreased by, for example, a given step size. Similarly, if the sign ofAccum_cmd is positive, the transmit power may be increased by, forexample, a given step size.

The reliability_threshold may be selected to correspond to a TPC commanderror rate of X %, for example. In this regard, a TPC command with anestimated reliability weight value corresponding to an error rate of X %or higher may be discarded from the calculation of the final TPC command512.

In another embodiment of the invention, the reliability weights wk maybe generated based on the TPC command signal and noise power estimatesfor each one of the plurality of radio link sets 1, . . . , k, asdescribed above with regard to FIG. 2. In this regard, the reliabilityweights wk may be determined from the following equation:$w_{k} = {{SNR}_{k} = \frac{{Stpc\_ avg}(k)}{{Ntpc\_ avg}(k)}}$where Stpc_avg(k) and Ntpc_avg(k) indicate the signal and noise power ofthe TPC command corresponding to RL set k.

Therefore, the overall accumulated command TPC_cmd 512 may be computedusing the following exemplary pseudo code:

Initialize the accumulated command to zero. Accum_cmd = 0For  (k = loop  over  RL  sets) $\begin{Bmatrix}{{Take}\quad{sign}\quad{of}\quad{TPC\_ cmd}(k)} \\{{Accum\_ cmd}+={\left( {{sign}\quad{of}\quad{TPC\_ sum}(k)} \right)*\frac{{Stpc\_ avg}(k)}{{Ntpc\_ avg}(k)}}}\end{Bmatrix}$

In another embodiment of the invention, in order to avoid computing theweights wk as a ratio, the TPC command signal and noise power estimatesmay be used in the determination of the total TPC command. The finalaccumulated command may then be determined by using the followingexemplary pseudo code:

Initialize the accumulated command to zero. Accum_cmd = 0For  (k = loop  over  RL  sets) $\begin{Bmatrix}{{Take}\quad{sign}\quad{of}\quad{TPC\_ cmd}(k)} \\{{Accum\_ cmd}+={\left( {{sign}\quad{of}\quad{TPC\_ sum}(k)} \right)*}} \\{{Stpc\_ avg}(k)*{\prod\limits_{j \neq k}{{Ntpc\_ avg}(j)}}}\end{Bmatrix}$

FIG. 6 is a flowchart illustrating exemplary steps for determining atotal TPC command in a WCDMA network, in accordance with an embodimentof the invention. Referring to FIGS. 5 and 6, at 602, the calculatedsignal power estimates (SPEs) for received TPC commands 502 a, . . . ,502 n from k RL sets may be received. At 604, the calculated noise powerestimates (NPEs) for the received TPC commands from k RL sets may bereceived. At 606, a reliability weight wk for each of the received TPCcommand for the k RL sets may be determined. At 608, the sign extractionblocks 504 a, . . . , 504 n may determine the sign for each received TPCcommand 502 a, . . . 502 n, respectively. At 610, a counter i may beincremented by 1. At 612, it may be determined whether reliabilityweight w_(i) is lower than a reliability threshold value. If thereliability weight w_(i) is lower than the reliability threshold value,at 614, w_(i) may be discarded from the calculation of the total TPCcommand 512. Processing may then resume at step 620. If the reliabilityweight w_(i) is greater than the reliability threshold value, at 616,the determined sign of the received TPC command for RL set i may bemultiplied by the corresponding reliability weight w_(i), to generateweighted TPC command w_TPC_(i). At 618, the total TPC command 512 may beincremented by the weighted TPC command w_TPC_(i). At 620, it may bedetermined whether i=k. If i is lower than k, processing may resume atstep 612. If i is equal to k, at 622, the transmit power adjustmentblock 514 may adjust transmit power based on the generated total TPCcommand 512.

In accordance with an embodiment of the invention, a method andapparatus for processing transmit power control (TPC) commands in awideband CDMA (WCDMA) network may comprise circuitry within the userequipment 120 that enables calculation of a signal-to-noise ratio (SNR)of a downlink dedicated physical channel (DPCH) 102 based on a pluralityof transmit power control (TPC) bits 156 received via the downlinkdedicated physical channel (DPCH) 102. A value of at least one of saidplurality of TPC bits 156 may not be known when said at least one ofsaid plurality of TPC bits 156 is received. The transmit poweradjustment block 514 within the user equipment 120 may enable adjustingof transmit power for at least one uplink communication path based onthe calculated SNR of the downlink dedicated physical channel 102. Atleast one processor within the user equipment 120, such as processor142, may enable calculation of at least one reliability weight value forat least a portion of the received TCP bits, based on the calculatedSNR.

The processor 142 within the user equipment 120 may enable generation ofa total TPC command for the at least one uplink communication path basedon the plurality of received TPC bits and the calculated at least onereliability weight value. The transmit power adjustment block 514 withinthe user equipment 120 may enable adjusting of the transmit power forthe at least one uplink communication path based on the generated totalTPC command. The processor 142 within the user equipment 120 may enablecalculation of the SNR based on a signal power of the DPCH 102 and/or anoise power of the DPCH 102. The summing block 206, 214, and/or 226within the user equipment 120 may enable summing of portions of theplurality of TPC bits that are received via a plurality of multipathsover the downlink dedicated physical channel to generate an in-phase (I)component and a quadrature (Q) component. The summing block 206, 214,and/or 226 within the user equipment 120 may enable summing of thegenerated I component and the generated Q component to determine signalpower of the DPCH.

The circuitry within the user equipment 120 may enable squaring of thesummed generated I component and the generated Q component to determinethe signal power of the DPCH. The circuitry within the user equipment120 may enable calculation of a norm of the squared summed I componentand generated Q component by dividing the squared summed I component andgenerated Q component by a number of the plurality of TPC bits per slotof the DPCH to determine the signal power of the DPCH. The averagingblock 222 or 232 within the user equipment 120 may enable averaging ofthe norm of the squared summed generated I component and generated Qcomponent over a time window. The processor 142 within the userequipment 120 may enable subtracting of the generated I component andthe generated Q component to determine noise power of the DPCH. Thecircuitry within the user equipment 120 may enable squaring of thesubtracted generated I component and the generated Q component todetermine the noise power of the DPCH.

The summing block 206, 214, and/or 226 within the user equipment 120 mayenable summing of the squared subtracted generated I component and thegenerated Q component over a plurality of TPC symbols to determine thenoise power of the DPCH 102. The processor 142 within the user equipment120 may enable calculation of a norm of the summed squared subtractedgenerated I component and the generated Q component by dividing thesummed squared subtracted generated I component and the generated Qcomponent by a number of the plurality of TPC bits per slot of the DPCHto determine the noise power of the DPCH 102. The processor 142 withinthe user equipment 120 may enable averaging of the norm of the summedsquared subtracted generated I component and the generated Q componentover a time window. The processor 142 within the user equipment 120 mayenable calculation of the SNR of the DPCH for a plurality of multipathsby averaging a calculated SNR of each of a plurality of radio link sets.

Another embodiment of the invention may provide a machine-readablestorage having stored thereon, a computer program having at least onecode section for signal processing, the at least one code section beingexecutable by a machine for causing the machine to perform steps asdisclosed herein.

Accordingly, aspects of the invention may be realized in hardware,software, firmware or a combination thereof. The invention may berealized in a centralized fashion in at least one computer system or ina distributed fashion where different elements are spread across severalinterconnected computer systems. Any kind of computer system or otherapparatus adapted for carrying out the methods described herein issuited. A typical combination of hardware, software and firmware may bea general-purpose computer system with a computer program that, whenbeing loaded and executed, controls the computer system such that itcarries out the methods described herein.

One embodiment of the present invention may be implemented as a boardlevel product, as a single chip, application specific integrated circuit(ASIC), or with varying levels integrated on a single chip with otherportions of the system as separate components. The degree of integrationof the system will primarily be determined by speed and costconsiderations. Because of the sophisticated nature of modernprocessors, it is possible to utilize a commercially availableprocessor, which may be implemented external to an ASIC implementationof the present system. Alternatively, if the processor is available asan ASIC core or logic block, then the commercially available processormay be implemented as part of an ASIC device with various functionsimplemented as firmware.

The invention may also be embedded in a computer program product, whichcomprises all the features enabling the implementation of the methodsdescribed herein, and which when loaded in a computer system is able tocarry out these methods. Computer program in the present context maymean, for example, any expression, in any language, code or notation, ofa set of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform. However, other meanings of computer program within theunderstanding of those skilled in the art are also contemplated by thepresent invention.

While the invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiments disclosed, but that the present inventionwill include all embodiments falling within the scope of the appendedclaims.

1. A method for signal processing, the method comprising: calculating asignal-to-noise ratio (SNR) of a downlink dedicated physical channel(DPCH) based on a plurality of transmit power control (TPC) bitsreceived via said downlink DPCH, wherein a value of at least one of saidplurality of TPC bits is not known when said at least one of saidplurality of TPC bits is received; and adjusting transmit power for atleast one uplink communication path based on said calculated SNR of saiddownlink dedicated physical channel.
 2. The method according to claim 1,further comprising calculating at least one reliability weight value forat least a portion of said received TCP bits, based on said calculatedSNR.
 3. The method according to claim 2, further comprising generating atotal TPC command for said at least one uplink communication path basedon said plurality of received TPC bits and said calculated at least onereliability weight value.
 4. The method according to claim 3, furthercomprising, if a selected one of said at least one reliability weightvalue is lower than a threshold value, discarding said selected one ofsaid at least one reliability weight value from said calculation of saidtotal TPC command.
 5. The method according to claim 4, wherein saidthreshold value is based on a TPC command error rate.
 6. The methodaccording to claim 3, further comprising adjusting said transmit powerfor said at least one uplink communication path based on said generatedtotal TPC command.
 7. The method according to claim 1, furthercomprising calculating said SNR based on at least one of the following:a signal power of said DPCH and a noise power of said DPCH.
 8. Themethod according to claim 1, further comprising summing portions of saidplurality of TPC bits that are received via a plurality of multipathsover said downlink dedicated physical channel to generate an in-phase(I) component and a quadrature (Q) component.
 9. The method according toclaim 8, further comprising: summing said generated I component and saidgenerated Q component to determine signal power of said DPCH; andsquaring said summed generated I component and said generated Qcomponent to determine said signal power of said DPCH.
 10. The methodaccording to claim 9, further comprising calculating a norm of saidsquared summed I component and said generated Q component by dividingsaid squared said summed I component and said generated Q component by anumber of said plurality of TPC bits per slot of said DPCH to determinesaid signal power of said DPCH.
 11. The method according to claim 10,further comprising averaging said norm of said I component and saidgenerated Q component over a time window.
 12. The method according toclaim 8, further comprising: subtracting said generated I component andsaid generated Q component to determine noise power of said DPCH; andsquaring said subtracted I component and said generated Q component todetermine said noise power of said DPCH.
 13. The method according toclaim 12, further comprising summing said squared I component and saidgenerated Q component over a plurality of TPC symbols to determine saidnoise power of said DPCH.
 14. The method according to claim 13, whereinsaid plurality of TPC symbols is half of said plurality of TPC bits perslot of said DPCH.
 15. The method according to claim 13, furthercomprising: calculating a norm of said summed I component and saidgenerated Q component by dividing said summed I component and saidgenerated Q component by a number of said plurality of TPC bits per slotof said DPCH to determine said noise power of said DPCH; and averagingsaid norm of said summed I component and said generated Q component overa time window.
 16. A system for signal processing, the systemcomprising: circuitry that enables calculation of a signal-to-noiseratio (SNR) of a downlink dedicated physical channel (DPCH) based on aplurality of transmit power control (TPC) bits received via saiddownlink DPCH, wherein a value of at least one of said plurality of TPCbits is not known when said at least one of said plurality of TPC bitsis received; and said circuitry enables adjusting of transmit power forat least one uplink communication path based on said calculated SNR ofsaid downlink dedicated physical channel.
 17. The system according toclaim 16, wherein said circuitry enables calculation of at least onereliability weight value for at least a portion of said received TCPbits, based on said calculated SNR.
 18. The system according to claim17, wherein said circuitry enables generation of a total TPC command forsaid at least one uplink communication path based on said plurality ofreceived TPC bits and said calculated at least one reliability weightvalue.
 19. The system according to claim 18, wherein said circuitryenables discarding of said selected one of said at least one reliabilityweight value from said calculation of said total TPC command, if aselected one of said at least one reliability weight value is lower thana threshold value.
 20. The system according to claim 19, wherein saidthreshold value is based on a TPC command error rate.
 21. The systemaccording to claim 18, wherein said circuitry enables adjusting of saidtransmit power for said at least one uplink communication path based onsaid generated total TPC command.
 22. The system according to claim 16,wherein said circuitry enables calculation of said SNR based on at leastone of the following: a signal power of said DPCH and a noise power ofsaid DPCH.
 23. The system according to claim 16, wherein said circuitryenables summing of portions of said plurality of TPC bits that arereceived via a plurality of multipaths over said downlink dedicatedphysical channel to generate an in-phase (I) component and a quadrature(Q) component.
 24. The system according to claim 23, wherein saidcircuitry enables summing of said generated I component and saidgenerated Q component to determine signal power of said DPCH, andwherein said circuitry enables squaring of said summed generated Icomponent and said generated Q component to determine said signal powerof said DPCH.
 25. The system according to claim 24, wherein saidcircuitry enables calculation of a norm of said squared summed Icomponent and said generated Q component by dividing said squared saidsummed I component and said generated Q component by a number of saidplurality of TPC bits per slot of said DPCH to determine said signalpower of said DPCH.
 26. The system according to claim 25, wherein saidcircuitry enables averaging of said norm of said I component and saidgenerated Q component over a time window.
 27. The system according toclaim 23, wherein said circuitry enables subtracting of said generated Icomponent and said generated Q component to determine noise power ofsaid DPCH, and wherein said circuitry enables squaring of saidsubtracted I component and said generated Q component to determine saidnoise power of said DPCH.
 28. The system according to claim 27, whereinsaid circuitry enables summing of said squared I component and saidgenerated Q component over a plurality of TPC symbols to determine saidnoise power of said DPCH.
 29. The system according to claim 28, whereinsaid plurality of TPC symbols is half of said plurality of TPC bits perslot of said DPCH.
 30. The system according to claim 28, wherein saidcircuitry enables calculation of a norm of said summed I component andsaid generated Q component by dividing said summed I component and saidgenerated Q component by a number of said plurality of TPC bits per slotof said DPCH to determine said noise power of said DPCH, and whereinsaid circuitry enables averaging of said norm of said summed I componentand said generated Q component over a time window.